Extremely high-speed switchmode DC-DC converters

ABSTRACT

Switchmode DC-DC power converters using one or more non-Silicon-based switching transistors and a Silicon-based (e.g. CMOS) controller are disclosed. The non-Silicon-based switching transistors may comprise, but are not necessarily limited to, III-V compound semiconductor devices such as gallium arsenide (GaAs) metal-semiconductor field effect transistors (MESFETs) or heterostructure FETs such as high electron mobility transistors (HEMTs). According to an embodiment of the invention, the low figure of merit (FoM), τ FET , of the non-Silicon-based switching transistors allows the converters of the present invention to be employed in envelope tracking amplifier circuits of wireless devices designed for high-bandwidth technologies such as, for example, EDGE and UMTS, thereby improving the efficiency and battery saving capabilities of the wireless devices.

FIELD OF THE INVENTION

The present invention relates generally to power conversion. Moreparticularly, the present invention relates to extremely high-speedswitchmode DC-DC converters.

BACKGROUND OF THE INVENTION

Switchmode DC-DC power converters are commonly used in electronicdevices. They operate by converting the voltage of an available voltagesource to a voltage that complies with the voltage requirements of oneor more components of the device. More particularly, switchmode DC-DCpower converters operate to convert a direct current (DC) input voltageto a DC output voltage that is higher (boost converter) or lower (buckconverter) than the DC input voltage. A common application of a buckconverter is the conversion of an available voltage source in a personalcomputer (PC) to a voltage compatible with the voltage sourcerequirements of devices and integrated circuits (e.g. the centralprocessing unit (CPU)) on the motherboard of the PC.

A conventional DC-DC buck converter 10 is shown in FIG. 1. A firstSilicon-based n-channel metal-oxide-semiconductor field effecttransistor (MOSFET) 12, commonly referred to as the “high-side switch”of the converter 10 is coupled to a second Silicon-based MOSFET 14,which is referred to as the “low-side switch”. Together, high-sideswitch 12 and low-side switch 14 control current flow through aninductor 16. During a charging phase of operation of converter 10, acontroller 18 maintains high-side switch 12 in an ON condition andmaintains low-side switch 14 in an OFF condition, thereby coupling a DCinput voltage, VIN, to inductor 16. This charging phase energizesinductor 16, which stores energy in its magnetic field. Following thecharging phase, converter 10 enters a discharging phase, during whichtime controller 18 maintains high-side switch 12 in an OFF condition andmaintains low-side switch 14 in an ON condition, thereby decoupling VINfrom inductor 16. During this discharging phase inductor 16 operates asa current source, supplying current from the energy stored in itsmagnetic field into capacitor 22 and load 24. Schottky diode 20 clampsany negative voltage from inductor 16 that may occur between theturn-OFF of switch 12 and the turn-ON of switch 14.

During the charging and discharging phases of operation of converter 10,current through inductor 16 rises and falls linearly, resulting in atriangular-shaped current signal. Capacitor 22 filters the inductorcurrent so that the output voltage, VOUT, of converter 10 is essentiallyDC. It can be shown that the average value of VOUT over time is equal tothe product of the duty cycle, D, of the high-side switch switchingperiod and the value of VIN. By way of a feedback loop 26, VOUT is fedback to controller 18, which dynamically compares VOUT to a referencevoltage VREF and modifies D depending on whether the value of VOUT ishigher than the desired output voltage level or lower than the desiredoutput voltage level.

In addition to switchmode power converters being of widespread use inthe PC market, they are also prevalent in the wireless device industry.In this technology sector, switchmode power converters are used to notonly provide efficient conversion for powering the baseband portion ofthe wireless device, but are also used to improve the efficiency of thepower amplifier (PA) of the radio frequency (RF) transmitter portion ofthe wireless device. (The PA is usually the dominant power consumer of awireless device.)

The PA of a wireless device is designed so that the battery voltagesupplied is large enough to permit maximum linear output voltage swingfor the largest RF signal present at the PA RF input. However, becausesmaller RF input levels (i.e. lower PA drive levels) require less DCpower for the same gain, the PA becomes inefficient at lower drivelevels. To improve efficiency at lower drive levels, a dynamic controltechnique known as “envelope tracking” has been developed. According tothis technique, the envelope of the PA RF input signal is tracked andused to regulate the battery voltage into a dynamically variable voltagesource. The envelope tracking technique thereby improves PA efficiency.When applied to a conventional linear amplifier this technique tends todegrade linearity, as it varies the bias of the active devices. However,when applied to polar transmitters there is no sacrifice of linearperformance, and the desired efficiency improvement is more readilyrealized.

Accurate envelope tracking requires that the switching frequency of theswitchmode DC-DC power converter be about 20 to 50 times higher than therequired signal envelope bandwidth. For a signal such as EDGE (EnhancedData GSM (Global System for Mobile Communications) Environment) thisenvelope bandwidth is 1 MHz, whereas for UMTS (Universal MobileTelecommunications System) this envelope bandwidth expands to 10 MHz.For EDGE, this means that the DC-DC power converter must switch at a20-50 MHz rate. This switching frequency requirement increases to200-500 MHz for UMTS application. Unfortunately, most DC-DC powerconverters operate with a switching frequency below 1 MHz, and a 2 MHzswitching frequency is considered to be extremely high. To meet thisefficiency need, therefore, there is a need to increase the switchingfrequency of DC-DC converters by a factor of about 20 to 200.

To evaluate the applicability of a variety of transistor types to thehigh-speed, high-current applications described above, a figure-of-merit(FoM) defined by the product of the transistor switch on-resistanceRDSon and the average input capacitance CGS of the transistor (definedas the ratio of the gate-charge QG required to turn on the FET to thegate-source voltage VGS required to set up the controlling electricfield, so that here CGS=QG/VGS) may be used. This FoM has units of timeand may be expressed as τ_(FET)=RDSon CGS. The validity of this approachis seen in FIG. 2, which plots several combinations of gate-charge andthe corresponding channel resistance achieved for Silicon MOSFETsavailable commercially. The best-fit τ_(FET) model is drawn among thepoints as a continuous curve. Different values of τ_(FET) are needed forthe PMOS and NMOS devices, which in this instance are 580 nanoseconds(ns) and 80 ns, respectively. It is also found that differentmanufacturers have slightly different values of τ_(FET) for theircompetitive devices, which demonstrates another intention for this FoMin that it should allow comparison of devices across process types.

Since the conducting channel of a FET is controlled by the electricfield between the gate and source terminals, the turning ON and OFF ofthis conducting channel depends on repeatedly moving this gate-chargeinto and out of the transistor. The value of the gate current requiredto move this charge depends on the amount of time allowed to move thecharge. Clearly, to achieve high operating speed it is strongly desiredto have a minimum amount of gate-charge necessary to control thechannel. Assuming that no more than 40% of the operating time is spentin switching transitions (a very generous assumption) then it ispossible to determine practical maximum values for τ_(FET), depending onthe desired operating frequency f_(CLK). This determination is presentedin FIG. 3. Commonly available values of τ_(FET) for modern SiliconMOSFET switching transistors range between 0.025 and 0.09 ns. As FIG. 3shows, these values limit the operating frequency of the Silicon MOSFETswitching transistor (using CMOS driver circuitry) to under 1 MHz. Toexceed this operating frequency it is necessary to use bipolar-baseddrive circuitry, a technique that is widely used in industry today. Thisis undesirable from both cost and integration compatibility points ofview. It is strongly desired to design using only CMOS technology forcost reasons. Any use of bipolar transistors on a chip forces the use ofa more expensive process. For most DC-DC switching converters the powerswitch transistors are external already, so process compatibility withthese huge transistors is not an issue anyway. How to control and drivethese huge transistors is an extremely important issue.

An alternative figure of merit, called FET-FOM, can be considered whichemphasizes the joint desirability of low switch on-resistance RDSon, lowgate-charge (QG), and low gate-source voltage (VGS). This is defined asthe product of these three parameters: FET-FOM=RDSon QG VGS. To comparevarious devices of different technologies, this alternative method hasmerit in that when gate-charge and gate-source voltage scale downtogether, the value of τ_(FET) will not change but the value of FET-FOMwill fall on the fact that both parameters are now lower.

To realize the desired efficiency improvements discussed above, theswitching transistors of the DC-DC power converter must be capable ofswitching ampere-scale currents within a small fraction of the period ofswitching frequency. For example, to use a switching frequency of 100MHz, the transistors must switch the supply currents ON or OFF intypically under one nanosecond. These requirements demand that thedriving circuitry in the controller of the converter be robust enough totranslate the low-level CMOS logic outputs of the controller 18 intodrive signals capable of driving switching transistors 12 and 14. Due tothe large gate capacitances of the switching transistors 12 and 14,however, the required size of the drivers could be prohibitively largeand in many instances, irrespective of size, simply unable to transferthe gate charge QG fast enough to switch the switching transistors atthe desired speed. This driver problem, in addition to the limits on theachievable τ_(FET)S of Silicon-based switching devices, renders theconventional converter 10 in FIG. 1 of no practical use for manyapplications including, for instance, use in the envelope trackingwireless device application described above.

There is prior art where GaAs technology is used to build the entireDC-DC converter, including GaAs device technology to drive the switchingtransistors of the converter. See, G. Hanington, A. Metzger, P. Asbeckand H. Finlay, “Integrated DC-DC Converter having GaAs HBT Technology”,Electronics Letters, vol. 35, pp. 2110-2112, 1999; M. Ranjan, K. H. Koo,C. Fallesen, G. Hanington and P. Asbeck, “Microwave Power Amplifierswith Digitally-Controlled Power Supply Voltage for High Efficiency andLinearity”, 2000 IEEE MIT-S International Microwave Symposium Digest,pp. 495, June 2000; S. Ajram, R. Kozlowski, H. Fawaz, D. Vandermoere andG. Salmer, “A fully GaAs-based 100 MHz, 2W DC-to-DC Power Converter”,Proceedings of the 27th European Solid-State Device Research Conference,Stuttgart, Germany, 22-24 September 1997. However, in addition to theseprior art approaches using all-GaAs technology (e.g. GaAs MESFETs and/orHBTs), all of the prior art in which HBTs are employed is applicable tothe boost (higher output voltage than input voltage) converterconfiguration only. That prior art is not applicable to the buck (loweroutput voltage than input voltage) converter configuration. HBT devices(or other bipolar devices) cannot be used for the shunt element(synchronous rectifier) because current flows in the reverse directionthrough these types of elements.

In summary, the gate-charge required by Silicon MOSFET transistors issimply too high to achieve the operating frequencies necessary tosupport EDGE, UMTS and other high-frequency envelope following DC-DCconverter applications. It would be desirable, therefore, to find analternative approach, which can both meet the desired operatingfrequencies and reduce the costs of driver circuitry by allowingstandard CMOS technology to be used.

SUMMARY OF THE INVENTION

Switchmode DC-DC power converters employing one or morenon-Silicon-based switching transistors and a Silicon-based (e.g. CMOS)controller are disclosed. The non-Silicon-based switching transistorsmay comprise, but are not necessarily limited to, III-V compoundsemiconductor devices such as gallium arsenide (GaAs)metal-semiconductor field effect transistors (MESFETs) orheterostructure FETs such as high electron mobility transistors (HEMTs).According to an embodiment of the invention, the low figure of merit(FoM), τ_(FET), of the non-Silicon-based switching transistors allowsthe converters of the present invention to be employed in envelopetracking amplifier circuits of wireless devices designed forhigh-bandwidth technologies such as, for example, EDGE and UMTS, therebyimproving the efficiency and battery saving capabilities of the wirelessdevices.

Further aspects of the invention are described and claimed below, and afurther understanding of the nature and advantages of the inventions maybe realized by reference to the remaining portions of the specificationand the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a conventional synchronous DC-DC buckconverter;

FIG. 2 is a graph illustrating the τ_(FET) characteristics of variousSilicon-based transistor switches of the prior art;

FIG. 3 is a graph illustrating the minimum τ_(FET) values required fordesired operating frequencies with direct drive from a standard CMOSdigital integrated circuit;

FIG. 4 is a schematic diagram of an exemplary DC-DC power converter,according to an embodiment of the present invention;

FIG. 5 is a schematic diagram of an alternative exemplary DC-DC powerconverter, according to an embodiment of the present invention;

FIG. 6 is a graph illustrating the τ_(FET) characteristics of aparticular exemplary switching transistor type, the GaAs MESFET, whichcan be used in the DC-DC power converters shown in FIG. 4, FIG. 5, orsimilar converter;

FIG. 7A is a table comparing FoM, i.e. τ_(FET), values of particularGaAs MESFET and pHEMT transistor switches to the τ_(FET) values ofvarious Silicon-based transistor switches of the prior art, when thetransistors are biased to their final gate-source voltages;

FIG. 7B is a graph comparing multiple transistor technologies using thealternative FET FoM;

FIG. 8 is a graph showing the ratio of a CMOS driver size to Si NMOSswitching transistor size as a function of switching transition time forcascaded CMOS gates made using transistors meeting the τ_(FET) valuesshown in FIG. 7A;

FIG. 9 is a graph relating the number of CMOS driver stages required vs.operating frequency for Si NMOS and GaAs pHEMT switching transistortypes;

FIG. 10 is a schematic block diagram of an envelope tracking amplifiercircuit, which may employ the DC-DC converter shown in FIG. 4, the DC-DCconverter shown in FIG. 5, or similar converter, so as to improve theefficiency of a power amplifier of a wireless device, according to anembodiment of the present invention;

FIG. 11 is a schematic block diagram of a direct polar transmittercircuit, which may employ the DC-DC converter shown in FIG. 4, the DC-DCconverter shown in FIG. 5, or similar converter, so as to improve theefficiency of a power amplifier of a wireless device, according to anembodiment of the present invention; and

FIG. 12 is a schematic block diagram of a high efficiency videoamplifier/driver circuit, which may employ the DC-DC converter shown inFIG. 4, the DC-DC converter shown in FIG. 5, or similar converter,according to an embodiment of the present invention.

DETAILED DESCRIPTION

Referring to FIG. 4, there is shown a schematic diagram of an exemplarysynchronous DC-DC power converter 40, according to an embodiment of thepresent invention. Converter 40 comprises a Silicon-based (e.g. CMOS)controller 41, non-Silicon-based switching transistors 42 and 44, aSchottky diode 46, an inductor 48, and a capacitor 49. FIG. 5 shows aschematic diagram of an alternative topology of a DC-DC converter 50,according to an alternative embodiment of the present invention.Converter 50 comprises a Silicon-based (e.g. CMOS) controller 52, anon-Silicon-based switching transistor 54, a diode 55, an inductor 56,and a capacitor 58. Unlike prior art power converters (e.g. the powerconverter 10 in FIG. 1), converters 40 and 50 are capable of followingwide-bandwidth envelope variations of wide-bandwidth technologies suchas, for example, EDGE and UMTS. Accordingly, as described in more detailbelow, the power converters of the present invention may be used to, forexample, improve the efficiencies and prolong the battery life ofwireless devices.

The ability to achieve the desired efficiencies and operatingfrequencies described above relates, at least in part, to the inventor'sdetermination that Silicon-based devices lack the necessary FoM, i.e.the τ_(FET), required to achieve such efficiencies and operatingfrequencies. FIG. 6 shows the τ_(FET) characteristics of a GaAs MESFETdevice, which is a transistor type that may be used for switchingtransistors 42 and 44 of converter 40 or for switching transistor 54 ofconverter 50 in the embodiments shown in FIGS. 4 and 5. When the gate ofthe GaAs MESFET is charged to its final gate-source voltage, the valueof τ_(FET) is approximately 0.0005 nanoseconds (ns). This τ_(FET) valueof the GaAs MESFET and the τ_(FET) value of a GaAs pseudomorphic highelectron mobility transistor (pHEMT) are compared to models of variousother Silicon transistor device technologies in FIG. 7A. From FIG. 7A itis seen that the GaAs MESFET has a τ_(FET) value that is a factor of 160times better (400 times better for the pHEMT) than the lowest τ_(FET)achieved by the other Silicon devices. This means that for a particulartransistor ON resistance (RDSon) required by the load application, theGaAs MESFET and pHEMT technologies exhibits a lower switch inputcapacitance and, therefore, require a lower amount of gate charge of theSilicon-based FET technologies evaluated in FIG. 7A. This distinction isfurther highlighted in FIG. 7B, where the alternative FET-FOM (RDSon QGVGS) is plotted for many instances of switching FETs of multipletechnologies. The Silicon-based devices all bunch together at the top,with values above unity. The GaAs MESFET devices have FET-FOM valuesapproximately two orders of magnitude lower, due primarily to thistechnology's much lower gate-charge requirements compared to Silicon.The GaAs pHEMT devices are yet another two orders of magnitude lower,due to their lower VGS voltages and slightly lower gate-charge. Clearlythe GaAs devices are far more desirable than the Silicon-basedtransistors for this high speed switch application.

GaAs EpHEMT devices have achieved operating frequencies of over 250 MHz.This is two orders of magnitude greater than frequencies consideredextremely high for all-Silicon switching transistor designs, placingthis approach well beyond evolutionary improvement status ofconventional designs. These results are achieved using pHEMT devicesthat are not specifically designed for this use, so further improvementsare expected as the inventions set forth in this disclosure arecapitalized upon by industry.

Those skilled in the art will understand that the particular GaAs MESFETand pHEMT devices described above are only exemplary and that othernon-Silicon-based device technologies using materials similar to orother than GaAs (e.g. other III/V compound semiconductors,high-temperature superconductors, etc.) may be used, so long as suchdevice technologies are capable of functioning as bi-directionalswitches and are able to exhibit superior τ_(FET) and/or gate-chargecharacteristics compared to the Silicon-based MOSFET switches describedin relation to the prior art or Similar τ_(FET) and/or gate-chargecharacteristics as that of the GaAs MESFET and pHEMT devices describedin the context of the various embodiments of the present invention.

Even if Silicon-based transistors could switch currents at the raterequired for a particular application (which as described above is notnecessarily the case), the driver circuitry required to switch theSilicon transistor would be large and costly, especially where thedriver circuitry comprises part of the Silicon-based controller of theconverter. The reason for this is that the gate capacitances ofSilicon-based MOSFETs are much larger than the gate capacitances ofnon-Silicon-based devices like the GaAs MESFET and pHEMT switchesdescribed above, for example.

When minimum transition time is the dominant performance metric, drivergates are sized to charge/discharge the load capacitance presented bythe power switching transistor. Large gate capacitances require smallerdriver on-resistances, with correspondingly larger driver devices, toachieve high switch transition speed. Therefore, faster switch operationrequires larger driver devices, for the same switching transistor. Ifthe speed requirement is fast enough, the driver devices may actually beno smaller than the actual power switch, and no gain is available toscale up from standard logic drive signals. Clearly this is a limitingcase of no value.

Consider, as an example, the drive requirements a one ampere switch madeusing GaAs MESFETs compared to the drive requirements of a switch madefrom Silicon-based MOSFETs. FIG. 8 shows that for cascaded CMOS gatesmade using Silicon transistors meeting the FoM shown in FIG. 7A, thisunity ‘gain’ point (where the switch driver transistor is the same sizeas the switch transistor itself) occurs at about 75 MHz operatingfrequency. FIG. 9 shows that when a Si NMOS switching transistor isused, the number of driver stages needed to attain this speed increasesdramatically as the operating frequency passes above 1/10 of this limit.(In this example, the power switch has an ON channel resistance of 0.05ohms, and the driving IC is limited to 30 ohms of drive resistance.These parameters are included not in any way to limit the scope of theinventions disclosed herein, but are included merely as a means ofhighlighting the difference in drive requirements between Silicon-basedswitching transistors and non-Si-based switching transistors.) Thesestages, while increasing in number, also increase in individual size,leading to an increase in controller die size, and ultimately anincrease in cost.

The power switch input capacitance is dramatically lower for GaAs MESFETand pHEMT switches than it is for a Silicon-based switches of equivalentcurrent capability. For example, the ratio between Silicon NMOS and GaAspHEMT input capacitances, found from FIG. 7A, is approximately0.08:0.0002˜400:1. The substantial drop in switch device equivalentinput capacitances renders the CMOS driver stages from the CMOS lineupin FIG. 9 no longer necessary. As shown in FIG. 9, the driver lineup forswitches made from GaAs pHEMT devices is essentially eliminated. Indeed,when the switching transistor is changed to a GaAs pHEMT device, theCMOS digital controller can readily drive the power switch alone, evenat a rate beyond 100 MHz. Accordingly, additional costs and Silicon areaare saved by the embodiments of the present invention. Avoiding use ofGaAs drive and control technology also saves costs and reducescomplexities. The cost of GaAs materials is much higher (on the order of5-10 times) than for Silicon. If GaAs is to be used, cost issues dictatethat a minimum amount of GaAs be allowed. Addressing these cost concernsand other performance demanding concerns, embodiments of the presentinvention allow only the power switches of the converter to be in anon-Silicon technology. All other circuitry, including driver circuitry,is implemented in Silicon technologies such as, for example, CMOS.

According to another embodiment of the invention, either of the DC-DCpower converters in FIGS. 4 and 5 (or similar power converter employingnon-Silicon-based switching transistors) may be used in an envelopetracking amplifier circuit of a wireless device transmitter to improvethe efficiency of the power amplifier (PA) of a radio frequency (RF)transmitter. An envelope tracking amplifier circuit 100 for improvingthe efficiency of a PA, according to these embodiments, is shown in FIG.10. An RF input signal is applied to an input port of an RF coupler 102,which operates to direct the RF input signal to an input port of a PA104 and couple the RF input signal to an input of an envelope detector106, as is known in the art. PA 104 produces an RF output signal that isradiated by an antenna 107. Envelope detector 106 operates to track theenvelope variation of the coupled RF input signal and provide a controlsignal to a control input of a DC-DC power converter 108, which asmentioned above may comprise either one of the power converters shown inFIG. 4 or 5 or similar power converter employing non-Silicon-basedswitching transistors. Based on the envelope variation, the controlsignal regulates a battery 110 into a dynamically variable voltage thatis used to power PA 104. Because non-Silicon-based, low τ_(FET) devicesare used in DC-DC power converter 108, envelope tracking amplifiercircuit 100 may be used to improve the efficiency and prolong thebattery life of wireless devices that use UMTS, EDGE and otherhigh-bandwidth technologies.

According to another embodiment of the invention, either of the DC-DCpower converters in FIGS. 4 and 5 (or similar power converter employingnon-Silicon-based switching transistors) may be used in a polar envelopemodulator circuit of a wireless device transmitter to improve theefficiency of the PA of the transmitter. A polar transmitter circuit 200for using an RF power generation stage (e.g. a PA) in its most efficientmanner, being fully compressed, according to these embodiments, is shownin FIG. 11. RF power generation stage 204 is a three port devicecomprising two input ports and one output port. A first input port 215is configured to receive an RF carrier signal with all angle modulation(if any). A second input port 217 is configured to receive an envelopecontrol signal from DC-DC converter 208. Modulation symbols are appliedto an input port of a polar modulator 205, which operates to convert themodulation symbols into their corresponding phase modulation (PM) andamplitude modulation (AM) components. The AM component is applied as acontrol signal to a control input of a DC-DC power converter 208, whichas mentioned above may comprise either one of the power converters shownin FIG. 4 or 5 or similar power converter employing non-Silicon-basedswitching transistors. RF power generation stage 204 produces an RFoutput signal at RF output port 219, which is then radiated by anantenna 207. Based on the amplitude modulation signal received frompolar modulator 205, the DC-DC converter 208 regulates a battery 210into a dynamically variable voltage that is used to power RF powergeneration stage 204. Because non-Silicon-based, low τ_(FET) devices areused in DC-DC power converter 208, polar transmitter circuit 200 may beused to provide long battery life for wireless devices that use UMTS,EDGE and other high-bandwidth technologies.

According to yet another embodiment of the invention, either of theDC-DC power converters in FIGS. 4 and 5 (or similar power converteremploying non-Silicon-based switching transistors) may be used as avideo driving amplifier to provide unusually high efficiency in such abroadband driver. A video amplifier/driver circuit 300 for providinghigh efficiency, according to these embodiments, is shown in FIG. 12. Avideo input signal is applied to both a non-inverting input port of anoperational amplifier 322 and a voltage summer 324. An offset voltage isadded to the video signal by the voltage summer 324, which provides acontrol signal to a control input of a DC-DC power converter 308, whichas mentioned above may comprise either one of the power converters shownin FIG. 4 or 5 or similar power converter employing non-Silicon-basedswitching transistors. Based on the offset video signal, the controlsignal regulates a battery 310 into a dynamically variable voltage thatis applied to pass transistor 320. The output terminal of passtransistor 320 is applied to the video load 330, shown in FIG. 12 as aresistor. The signal on the output terminal of pass transistor 320 isalso fed back to the inverting input of operational amplifier 322. Theoutput of operational amplifier 322 is connected to the control terminalof pass transistor 320. Because non-Silicon-based, low τ_(FET) devicesare used in DC-DC power converter 108, envelope tracking amplifiercircuit 100 may be used to improve the efficiency of video drivingamplifiers, thereby reducing the heat they dissipate.

Whereas the above is a complete description of the preferred embodimentsof the invention, various alternatives, modifications, and equivalentsmay be used. Therefore, the above description should not be taken aslimiting the scope of the invention as it is defined by the appendedclaims.

1-57. (canceled)
 58. A switchmode DC-DC converter, comprising: a firstnon-Silicon-based switching transistor having a drain configured toconnect to a DC input voltage, a source and a gate; a Silicon-basedcontroller having an output coupled to the gate of the firstnon-Silicon-based switching transistor; an inductor having a first endcoupled to the source of the first non-Silicon-based switchingtransistor and a second end embodying an output of the converter; and adiode having a first end coupled to the source of the firstnon-Silicon-based switching transistor and to the first end of theinductor. 3